Medical device

ABSTRACT

A medical device is described having a handle, a shaft coupled to the handle and an end effector coupled to the shaft. In one embodiment, the device includes an ultrasonic transducer and is arranged so that ultrasonic or electrical energy can be delivered to a vessel or tissue to be treated. Various novel sensing circuits are described to allow a measure of the drive signal to be measured and fed back to a controller. An active fuse circuit is also described for protecting one or more batteries of the device from an over-current situation.

The present invention relates to the field of medical devices and inparticular, although not exclusively, to medical cauterization andcutting devices. The invention also relates to drive circuits andmethods for driving such medical devices.

Many surgical procedures require cutting or ligating blood vessels orother internal tissue and many procedures are performed using minimallyinvasive techniques with a hand-held cauterization device to perform thecutting or ligating. Some existing hand-held cauterization devices usean ultrasonic transducer in the cauterization device to apply ultrasonicenergy to the tissue to be cut or ligated. Other hand-held cauterizationdevices apply RF energy directly to the tissue/vessel being cauterizedvia forceps of the device.

The present invention aims to provide an alterative surgical device thatis able to apply ultrasonic energy or RF energy to the vessel or tissueto be cauterized. Other aspects of the invention relate to the way inwhich control circuitry is provided to select between the differentoperating modes. Other aspects of the invention relate to the way inwhich voltage and current measurements can be made in the circuit designfor reporting to a controller, such as a microprocessor; and to the wayin which control circuitry can be provided to ensure that too muchcurrent is not drawn from the battery.

According to one aspect, the present invention provides a medical devicecomprising: an end effector for gripping a vessel/tissue; an ultrasonictransducer coupled to the end effector; a drive circuit coupled to theend effector and to the ultrasonic transducer and operable to generate aperiodic drive signal and to provide the drive signal either to theultrasonic transducer or to the end effector; and a controller operableto control the drive circuit so that the drive signal is applied to adesired one of the ultrasonic transducer and the end effector.

In one embodiment, the drive circuit comprises a first resonant circuithaving a first resonant frequency and a second resonant circuit having asecond resonant frequency that is different to the first resonantfrequency, wherein the first resonant frequency corresponds to aresonant characteristic of the ultrasonic transducer and wherein thecontroller is operable to control the drive circuit so that the drivecircuit generates a drive signal having a frequency corresponding to thefirst resonant frequency when the drive signal is to be applied to theultrasonic transducer and so that the drive circuit generates a drivesignal having a frequency corresponding to the second resonant frequencywhen the drive signal is to be applied to the end effector.

A signal generator may also be provided that is coupled between thecontroller and the drive circuit for generating a cyclically varyingvoltage from a DC voltage supply in dependence upon control signals fromthe controller and for supplying the cyclically varying voltage to thefirst and second resonant circuits of the drive circuit.

The controller may be arranged to vary the period of the drive signalabout the first resonant frequency or the second resonant frequency tovary the energy supplied to the vessel or tissue gripped by the endeffector. The controller may vary the period of the drive signal so thatthe frequency of the drive signal varies around the first resonantfrequency within 0.1% to 1% of the first resonant frequency; or so thatthe frequency of the drive signal varies around the second resonantfrequency within 40% to 60% of the second resonant frequency.

Typically, the resonant characteristics of the first and second resonantcircuits vary with the tissue or vessel gripped by the forceps and inone embodiment, the controller is configured to vary the period of thedrive signal to track changes in the respective resonant characteristic.

An ultrasonic waveguide may be provided that is coupled to theultrasonic transducer for guiding ultrasonic energy generated by theultrasonic transducer towards the end effector. The end effector maycomprise first and second jaws and the second resonant circuit may beelectrically coupled to the first and second jaws of the end effector.For example, the first jaw of the end effector may be electricallycoupled to the waveguide and the second resonant circuit may beelectrically coupled to the first jaw of the end effector via theultrasonic waveguide. In some embodiments, the first resonant circuit iselectrically coupled to the ultrasonic transducer and to the waveguide.

Sensing circuitry may be provided for sensing a drive signal applied tothe ultrasonic transducer or to the end effector. In one embodiment, oneor both of the first and second resonant circuits may comprise at leastone of an inductor coil, a capacitor and a resistor and wherein thesensing circuitry may comprise an op-amp circuit for sensing the voltageacross the inductor coil or the capacitor or the resistor and forconverting the sensed voltage to a sensor signal suitable for inputtingto the controller. In an alternative embodiment, one or both of thefirst and second resonant circuits may comprise an impedance elementthat is connected between the resonant circuit and a reference potentialand wherein the sensing circuitry comprises a divider circuit forobtaining a measure of the voltage across the impedance element and abias signal generator for applying a DC bias signal to the voltagemeasure. In this case, the impedance element may comprise a capacitor ora resistor. Typically, the sensing circuitry comprises DC blockingcircuitry for preventing the DC bias signal from the bias signalgenerator from coupling with the drive circuit. The bias signalgenerator may comprises a voltage divider circuit connected between areference voltage and a supply voltage of the controller.

The device is preferably a battery operated device and comprises one ormore batteries for powering the device and further comprising an activefuse circuit for protecting the one or more batteries. The active fusecircuit may comprise a switch electrically coupled between a terminal ofthe one or more batteries and the drive circuit and control circuitryconfigured to open the switch to isolate the supply terminal from thedrive circuit.

The present invention also provides a medical device comprising: an endeffector for gripping a vessel/tissue; a drive circuit for generating acyclically varying drive signal for driving energy into thevessel/tissue; sensing circuitry for sensing a drive signal generated bythe drive circuit; and a controller responsive to the sensing circuitryand operable to control the drive circuit to control the energydelivered to the vessel/tissue; wherein the drive circuit comprises animpedance element that is coupled to a reference potential and whereinthe sensing circuitry comprises a divider circuit for obtaining ameasure of the voltage across the impedance element and a bias signalgenerator for applying a DC bias signal to the voltage measure.

The impedance element may be a capacitor or a resistor. The sensingcircuitry may also comprise DC blocking circuitry for preventing the DCbias signal from the bias signal generator from coupling with the drivecircuit. The bias signal generator may comprise a divider circuitconnected between a reference voltage and a supply voltage of thecontroller. The divider circuit of the bias signal generator may beconnected to the supply voltage of the controller via a switch andwherein the controller is configured to open the switch when thecontroller does not require signals from the sensing circuitry.

The present invention also provides a medical device comprising: an endeffector for gripping a vessel/tissue; one or more batteries forproviding a DC voltage supply; a signal generator coupled to the one ormore batteries for generating a cyclically varying drive signal from theDC voltage supply for driving energy into the vessel/tissue; acontroller operable to control the signal generator to control theenergy delivered to the vessel/tissue; and an active fuse circuitcoupled between the one or more batteries and the signal generator forprotecting the one or more batteries.

The active fuse circuit may comprise a switch that is electricallycoupled between a terminal of the one or more batteries and the signalgenerator; and control circuitry configured to switch the switch. Theswitch may be arranged to disconnect the signal generator from the oneor more batteries or may connect a large impedance between the signalgenerator and the one or more batteries.

The control circuitry of the active fuse may comprise circuitry forsensing a measure of the current being drawn from the one or morebatteries and is configured to switch the switch in the event that thecurrent measure exceeds a threshold. The control circuitry of the activefuse may comprise a comparator for comparing the current measure withthe threshold and wherein an output of the comparator controls theopening and closing of the switch.

The present invention also provides a method of operating a medicaldevice comprising generating a periodic drive signal and applying thedrive signal to an ultrasonic transducer or to an end effector of themedical device and controlling the drive circuit so that the drivesignal is applied to a desired one of the ultrasonic transducer and theend effector.

The present invention also provides a method of cauterising or cutting avessel or tissue, the method comprising: gripping the vessel or tissuewith an end effector of a medical device; using a drive circuit to applya periodic drive signal either to an ultrasonic transducer or to the endeffector; and controlling the drive circuit so that the drive signal isapplied to a desired one of the ultrasonic transducer and the endeffector. The method may use the above described medical device.

The present invention also provides electronic apparatus for use in amedical device having an ultrasonic transducer and an end effector, theelectronic apparatus comprising: a drive circuit for generating aperiodic drive signal; and a controller operable to control the drivecircuit so that the drive signal is applied to a desired one of theultrasonic transducer and the end effector; wherein the drive circuitcomprise a first resonant circuit having a first resonant frequency anda second resonant circuit having a second resonant frequency that isdifferent to the first resonant frequency, and wherein the controller isoperable to control the drive circuit so that the drive circuitgenerates a drive signal having a frequency corresponding to the firstresonant frequency when the drive signal is to be applied to theultrasonic transducer and so that the drive circuit generates a drivesignal having a frequency corresponding to the second resonant frequencywhen the drive signal is to be applied to the end effector.

The present invention also provides a medical device comprising: an endeffector for gripping a vessel/tissue; a drive circuit coupled to theend effector and operable to generate a drive signal and to provide thedrive signal to the end effector; a controller operable to generate andoutput control signals to the drive circuit to control the drive signalgenerated by the drive circuit; wherein the drive circuit and a loadformed by the vessel/tissue gripped by the end effector define aresonant circuit whose resonant frequency varies as the impedance of theload formed by the vessel/tissue gripped by the end effector changes;wherein the controller is arranged to generate control signals whichcause the drive circuit to generate a drive signal having a frequencythat tracks said resonant frequency as it changes; and wherein saidcontroller is further arranged to reduce one or more of the power,current or voltage delivered to the load formed by the vessel/tissuegripped by the end effector.

Sensor circuitry may be provided for sensing signals applied to the loadformed by the vessel/tissue gripped by the end effector and measurementcircuitry for processing the signals from the sensor circuitry todetermine a measure of the impedance of the load formed by thevessel/tissue gripped by the end effector. In this case, the controllercan generate said control signals in dependence upon said measure of theimpedance of the load formed by the vessel/tissue gripped by the endeffector.

In one embodiment, the controller generates control signals havingsequences of pulses and the controller skips one or more pulses from thecontrol signals in order to reduce one or more of the power, current orvoltage delivered to the load formed by the vessel/tissue gripped by theend effector.

Typically, in this case, the controller comprises a pulse signalgenerator that generates pulses at a desired frequency that depends onsaid resonant frequency and the controller skips pulses generated bysaid pulse signal generator by suppressing pulses generated by the pulsesignal generator.

These and various other features and aspects of the invention willbecome apparent from the following detailed description of embodimentswhich are described with reference to the accompanying Figures in which:

FIG. 1 illustrates a hand-held cauterization device that has batteriesand drive and control circuitry mounted in a handle portion of thedevice;

FIG. 2 is a part block diagram illustrating the main components of thecauterization device used in one embodiment of the invention;

FIG. 3 is a circuit diagram illustrating the main electrical componentsof the cauterization device shown in FIG. 2;

FIG. 4 schematically illustrates the way in which the ultrasonictransducer is coupled to a waveguide for delivering the generatedultrasonic energy to the forceps and illustrating the way in which thecircuitry shown in FIG. 3 can deliver electrical energy to the forceps;

FIG. 5 is a block diagram that schematically illustrates processingmodules that form part of the microprocessor shown in FIG. 2;

FIG. 6 illustrates the form of control signals generated by themicroprocessor to control the drive circuit whilst minimising 3^(rd)harmonic content;

FIG. 7 is a contour plot illustrating the delivered power versus theload resistance and the drive frequency;

FIG. 8a is a circuit diagram illustrating one way in which a measure ofthe load current can be determined and supplied to the microprocessor;

FIG. 8b is a circuit diagram illustrating another way in which a measureof the load current can be determined and supplied to themicroprocessor;

FIG. 8c is a circuit diagram illustrating one way in which a measure ofthe load voltage can be determined and supplied to the microprocessor;

FIG. 9a is a circuit diagram illustrating one way in which a measure ofthe load current can be determined and supplied to the microprocessorwithout using an op-amp circuit;

FIG. 9b is a circuit diagram illustrating one way in which a measure ofthe load current can be determined and supplied to the microprocessorand illustrating one way in which a measure of the load voltage can bedetermined and supplied to the microprocessor without using op-ampcircuits; and

FIG. 10 is a circuit diagram illustrating an active fuse circuit used toprotect the batteries shown in FIG. 2 from excessive current demand.

MEDICAL DEVICE

Many surgical procedures require cutting or ligating blood vessels orother vascular tissue. With minimally invasive surgery, surgeons performsurgical operations through a small incision in the patient's body. As aresult of the limited space, surgeons often have difficulty controllingbleeding by clamping and/or tying-off transected blood vessels. Byutilizing ultrasonic-surgical forceps or electro-surgical forceps, asurgeon can cauterize, coagulate/desiccate, and/or simply reducebleeding by controlling the ultrasonic energy applied to thetissue/vessel by an ultrasonic transducer or by controlling the RFenergy applied to the tissue/vessel via the forceps.

FIG. 1 illustrates the form of an ultrasonic/RF-surgical medical device1 that is designed for minimally invasive medical procedures, accordingto one embodiment of the present invention. As shown, the device 1 is aself contained device, having an elongate shaft 3 that has a handle 5connected to the proximal end of the shaft 3 and an end effector 7connected to the distal end of the shaft 3. In this embodiment, the endeffector 7 comprises medical forceps 9 that are controlled by the usermanipulating control levers 11 and 13 of the handle 5.

During a surgical procedure, the shaft 3 is inserted through a trocar togain access to the patient's interior and the operating site. Thesurgeon will manipulate the forceps 9 using the handle 5 and the controllevers 11 and 13 until the forceps 9 are located around the vessel to becut or cauterised. Electrical energy is then applied, in a controlledmanner, either to the tissue directly via the forceps 9 (as RF energy)or to an ultrasonic transducer 8 that is mounted within the handle 5 andcoupled to the forceps 9 via a waveguide (not shown) within the shaft 3,in order to perform the desired cutting/cauterisation using ultrasonicenergy. As shown in FIG. 1, in this embodiment, the handle 5 also housesbatteries 15 and control electronics 17 for generating and controllingthe electrical energy required to perform the cauterisation. In thisway, the device 1 is self contained in the sense that it does not need aseparate control box and supply wire to provide the electrical energy tothe forceps 9. However, such a separate control box may be provided ifdesired.

System Circuitry

FIG. 2 is a schematic block diagram illustrating the main electricalcircuitry of the cauterization/cutting device 1 used in this embodimentto generate and control the electrical energy supplied to the ultrasonictransducer or to the forceps 9. As will be explained in more detailbelow, in this embodiment, the circuitry is designed to control theperiod of an electrical drive waveform that is generated in order tocontrol the amount of power delivered to the tissue/vessel beingcauterized.

As shown in FIG. 2, the cauterization/cutting device 1 comprises a userinterface 21—via which the user is provided with information (such as anindication that energy is being applied to the gripped tissue/vessel byelectrical energy or ultrasonic energy) and through which the usercontrols the operation of the cauterization/cutting device 1, includingselection of ultrasonic operation or RF operation. As shown, the userinterface 21 is coupled to a microprocessor 23 that controls thecutting/cauterisation procedure by generating control signals that itoutputs to gate drive circuitry 25. In response to the control signalsfrom the microprocessor 23, the gate drive circuitry 25 generates gatecontrol signals that cause a bridge signal generator 27 to generate adesired drive waveform that is applied either to the ultrasonictransducer 8 or to the forceps 9 via a drive circuit 29. Voltage sensingcircuitry 31 and current sensing circuitry 33 generate measures of thecurrent and voltage applied to the ultrasonic transducer 8 or to theforceps 9, which they feed back to the microprocessor 23 for controlpurposes. FIG. 2 also shows the batteries 15 that provide the power forpowering the electrical circuitry shown in FIG. 2. In this embodiment,the batteries 15 are arranged to supply 0V and 14V rails.

FIG. 3 illustrates in more detail the components of the gate drivecircuitry 25, the bridge signal generator 27 and the drive circuit 29.FIG. 3 also shows an electrical equivalent circuit 30 of thepiezo-electric ultrasonic transducer 8 and the load (R_(load)) formed bythe tissue/vessel to be treated. As shown in FIG. 3, the gate drivecircuitry 25 includes two FET gate drives 37—FET gate drive 37-1 and FETgate drive 37-2. A first set of control signals (CTRL₁) from themicroprocessor 23 is supplied to FET gate drive 37-1 and a second set ofcontrol signals (CTRL₂) from the microprocessor 23 is supplied to FETgate drive 37-2. FET gate drive 37-1 uses the first set of controlsignals (CTRL₁) to generate two drive signals—one for driving each ofthe two FETs 41-1 and 41-2 of the bridge signal generator 27. The FETgate drive 37-1 generates drive signals that causes the upper FET (41-1)to be on when the lower FET (41-2) is off and vice versa. This causesthe node A to be alternately connected to the 14V rail (when FET 41-1 isswitched on) and the 0V rail (when the FET 41-2 is switched on).Similarly, FET gate drive 37-2 uses the second set of control signals(CTRL₂) to generate two drive signals—one for driving each of the twoFETs 41-3 and 41-4 of the bridge signal generator 27. The FET gate drive37-2 generates drive signals that causes the upper FET (41-3) to be onwhen the lower FET (41-4) is off and vice versa. This causes the node Bto be alternately connected to the 14V rail (when FET 41-3 is switchedon) and the 0V rail (when the FET 41-4 is switched on). Thus the twosets of control signals (CTRL₁ and CTRL₂) output by the microprocessor23 control the digital waveform that is generated and applied betweennodes A and B. Each set of control signals (CTRL₁ and CTRL₂) comprisesof a pair of signal lines, one to indicate when the high side FET is onand the other to indicate when the low side FET is on. Thus themicroprocessor 23, either through software or through a dedicatedhardware function can ensure that the undesirable condition when bothhigh and low side FETs are simultaneously turned on does not occur. Inpractice this requires leaving a dead time when both high and low sideFETs are turned off to ensure that, even when allowing for variableswitching delays, there is no possibility that both FETs can besimultaneously on. In the present embodiment a dead time of about 100 nswas used.

As shown in FIG. 3, the nodes A and B are connected to the drive circuit29, thus the digital voltage generated by the bridge signal generator 27is applied to the drive circuit 29. This applied voltage will causecurrent to flow in the drive circuit 29. As shown in FIG. 3, the drivecircuit 29 includes two transformer circuits 42-1 and 42-2. The firsttransformer circuit 42-1 is designed for efficient driving of theultrasonic transducer 8 and includes a capacitor-inductor-inductorresonant circuit 43-1 formed by capacitor C^(US) _(s) 45, inductorL^(US) _(s) 47 and inductor L^(US) _(m) 49. When driving the ultrasonictransducer 8, the microprocessor 23 is arranged to generate controlsignals for the gate drive circuitry 25 so that the fundamentalfrequency (f_(d)) of the digital voltage applied across nodes A and B isaround the resonant frequency of the resonant circuit 43-1, which inthis embodiment is about 50 kHz. As a result of the resonantcharacteristic of the resonant circuit 43-1, the digital voltage appliedacross nodes A and B will cause a substantially sinusoidal current atthe fundamental frequency (f_(d)) to flow within the resonant circuit43-1. This is because higher harmonic content of the drive voltage willbe attenuated by the resonant circuit 43-1 and the impedance of L_(t)and C_(t1) referred to the transformer primary.

As illustrated in FIG. 3, the inductor L^(US) _(m) 49 forms the primaryof the transformer circuit 42-1, the secondary of which is formed byinductor L^(US) _(sec) 53. The transformer up-converts the drive voltage(V^(US) _(d)) across the inductor L_(m) 49 to a load voltage (V_(L);typically about 120 volts) that is applied to the ultrasonic transducer8. The electrical characteristics of the ultrasonic transducer 8 changewith the impedance of the forceps' jaws and any tissue or vessel grippedby the forceps 9; and FIG. 3 models the ultrasonic transducer 8 and theimpedance of the forceps' jaws and any tissue or vessel gripped by theforceps 9 by the inductor L_(t) 57, the parallel capacitors C_(t1) 59and C_(t2) 61 and the resistance R_(load).

The inductor L^(US) _(s) and capacitor C^(US) _(s) of the drive circuit29 are designed to have a matching LC product to that of inductor L_(t)and capacitor C_(t1) of the ultrasonic transducer 8. Matching the LCproduct of a series LC network ensures that the resonant frequency ofthe network is maintained. Similarly, the magnetic reactance of theinductor L^(US) _(m) is chosen so that at resonance it matches with thecapacitive reactance of the capacitor C_(t2) of the ultrasonictransducer 8. For example, if the transducer 8 is defined such thatcapacitor C_(t2) has a capacitance of about 3.3 nF, then the inductorL^(US) _(m) should have an inductance of about 3 mH (at a resonantfrequency of about 50 kHz). Designing the drive circuit 29 in this wayprovides for the optimum drive efficiency in terms of energy delivery tothe tissue/vessel gripped by the forceps 9. The efficiency improvementis realised because the current flowing in C^(US) _(s) and consequentlythe FET bridge (27) is reduced, because the transformer magnetisingcurrent cancels out the current flowing in C_(t2) In addition, becauseof this current cancellation, the current flowing in C^(US) _(s) isproportional to the current flowing in Rload, which allows the loadcurrent to be determined by measuring the current flowing in C^(US)_(s).

The second transformer circuit 42-2 is designed for efficient driving ofelectrical RF energy directly to the tissue/vessel via the forceps 9 andincludes a capacitor-inductor-inductor resonant circuit 43-2 formed bycapacitor C^(F) _(s) 46, inductor L^(F) _(s) 48 and inductor L^(F) _(m)50. When driving the forceps 9 directly with electrical energy, themicroprocessor 23 is arranged to generate control signals for the gatedrive circuitry 25 so that the fundamental frequency (f_(d)) of thedigital voltage applied across nodes A and B is around the resonantfrequency of the resonant circuit 43-2, which in this embodiment isabout 500 kHz. As a result of the resonant characteristic of theresonant circuit 43-2, the digital voltage applied across nodes A and Bwill cause a substantially sinusoidal current at the fundamentalfrequency (f_(d)) to flow within the resonant circuit 43-2. This isbecause higher harmonic content of the drive voltage will be attenuatedby the resonant circuit 43-2.

As illustrated in FIG. 3, the inductor L^(F) _(m) 50 forms the primaryof the transformer circuit 42-1, the secondary of which is formed byinductor L^(US) _(sec) 54. The transformer up-converts the drive voltage(V^(F) _(d)) across inductor L^(F) _(m) 50 to the load voltage (V^(F)_(L); typically about 120 volts) that is applied to the forceps 9. Thetissue or vessel gripped by the jaws of the forceps 9 is represented asthe resistive load R_(load) in the box labelled 9 in FIG. 3. Inpractice, this will be the same resistive load that is illustrated inthe electrical equivalent circuit 30 of the ultrasonic transducer 8.

FIG. 4 is a schematic diagram illustrating the way in which theultrasonic transducer 8 couples to the tissue/vessel to be cauterizedand the way in which the circuit components illustrated in FIG. 3connect to the ultrasonic transducer 8 and to the forceps 9. Inparticular, FIG. 4 shows the shaft 3, the forceps 9 and the ultrasonictransducer 8. FIG. 4 also shows the waveguide 72 along which theultrasonic signal that is generated by the ultrasonic transducer 8 isguided. The waveguide 72 is connected to the node “BB” show in FIG. 3,whilst the input supply to the ultrasonic transducer 8 is connected tonode “AA” shown in FIG. 3. The output node “CC” of the secondtransformer circuit 42-2 is connected to a conductive inner wall of thesheath 3, which is electrically connected to the upper jaw 74 of theforceps 9. The return path is through the tissue/vessel to be cauterizedand the lower jaw 76, which is electrically connected to the node “BB”.

When the drive signal has a drive frequency of about 50 kHz, very littlecurrent will flow within the second transformer circuit 42-2 because thedrive frequency is far away from the resonant frequency of the resonantcircuit 43-2 such that the input impedance of the second transformercircuit 42-2 will be very high for this drive signal. Therefore, thepower will be delivered almost entirely via the first transformercircuit 42-1. Similarly, when the drive signal has a drive frequency ofabout 500 kHz, very little current will flow within the firsttransformer circuit 42-1 because the drive frequency is far away fromthe resonant frequency of the resonant circuit 43-1 such that the inputimpedance of the first transformer circuit 42-1 will be very high forthis drive signal. Therefore, the power will be delivered almostentirely via the second transformer circuit 42-2. In this way, the twotransformer circuits 42-1 and 42-2 can be driven by a common bridgesignal generator 27; although it is also feasible to drive eachtransformer circuit with separate bridge signal generators.

It is not always desired to apply full power to the tissue/vessel to betreated. Therefore, in this embodiment in the ultrasonic mode ofoperation, the amount of ultrasonic energy supplied to the vessel/tissueis controlled by varying the period of the digital waveform appliedacross nodes A and B so that the drive frequency (f_(d)) moves away fromthe resonant frequency of the ultrasonic transducer 8. This worksbecause the ultrasonic transducer 8 acts as a frequency dependent(lossless) attenuator. The closer the drive signal is to the resonantfrequency of the ultrasonic transducer 8, the more ultrasonic energy theultrasonic transducer 8 will generate. Similarly, as the frequency ofthe drive signal is moved away from the resonant frequency of theultrasonic transducer 8, less and less ultrasonic energy is generated bythe ultrasonic transducer 8. In addition or instead, the duration of thepulses of the drive signals may be varied to control the amount ofultrasonic energy delivered to the tissue/vessel.

Similarly, in the electrical mode of operation, the amount of electricalpower supplied to the forceps 9 is controlled by varying the period ofthe digital waveform applied across nodes A and B so that the drivefrequency (f_(d)) moves away from the resonant frequency of the resonantcircuit 43-2. This works because the resonant circuit 43-1 acts as afrequency dependent (lossless) attenuator. The closer the drive signalis to the resonant frequency of the resonant circuit 43-1, the less thedrive signal is attenuated. Conversely, as the frequency of the drivesignal is moved away from the resonant frequency of the circuit 43-1,the more the drive signal is attenuated and so the electrical energysupplied to the tissue/vessel reduces. The drive frequency needs to moveaway from the resonant frequency by about 50% of the resonant frequencyto achieve the desired range of power variation. An alternative approachto controlling the power, current or voltage applied during theelectrical mode of operation is to continuously tune the frequency ofthe excitation signal to keep it matched with the resonant frequency ofthe drive circuit (as it changes with the changing load impedance) andthereby maintain efficient operation and to skip some of the pulses ofthe drive control signals until the average power, current and/orvoltage is below the relevant limit.

A further alternative, which is most effective when using the pulseskipping technique, is to remove the inductor 48 shown in FIG. 3 andtherefore the drive circuit for the electrical mode of operation becomesa substantially parallel LC resonant circuit (it is not a pure parallelLC resonant circuit because the transformer leakage inductance appearsin series with inductor 48 and cannot be entirely removed). Theadvantage of removing the inductor 48 is that the overall efficiency canbe increased, because there are no longer any losses in the inductor. Afurther advantage is that the physical size of the circuit can bereduced, because often the inductor is a physically large componentrelative to the FETs, microprocessor, capacitors and other systemcomponents.

The microprocessor 23 controls the power delivery based on a desiredpower to be delivered to the circuitry 30 (which models the ultrasonictransducer 8 and the tissue/vessel gripped by the forceps 9) or to theforceps 9 and based on measurements of the load voltage (V_(L)) and ofthe load current (i_(L)) obtained from the voltage sensing circuitry 31and the current sensing circuitry 33. The microprocessor 23 also selectsthe frequency of the drive signal (around 50 kHz or around 500 kHz)based on a user input received via the user interface 21 that selectseither electrical operation or ultrasonic operation.

Microprocessor

FIG. 5 is a block diagram illustrating the main components of themicroprocessor 23 that is used in this embodiment. As shown, themicroprocessor 23 includes synchronous I,Q sampling circuitry 81 thatreceives the sensed voltage and current signals from the sensingcircuitry 31 and 33 and obtains corresponding samples which are passedto a measured voltage and current processing module 83. The measuredvoltage and current processing module 83 uses the received samples tocalculate the impedance of, and the RMS voltage applied to and the RMScurrent flowing through, the ultrasonic transducer 8 and/or directly tothe tissue/vessel gripped by the forceps 9; and from them the power thatis presently being supplied to the circuitry 30 or directly to thetissue/vessel gripped by the forceps 9. The determined values are thenpassed to a power controller 85 for further processing. The measuredvoltage and current processing module 83 can also process the received Iand Q samples to calculate the phase difference between the load voltage(V_(L)) and the load current (i_(L)). During the ultrasonic mode ofoperation, at resonance, this phase difference should be around zero andso this phase measure can be used as a feedback parameter for the powercontroller 85.

The power controller 85 uses the received impedance value and thedelivered power value to determine, in accordance with a predefinedalgorithm and a power set point value and a mode indication signal(received from a medical device control module 89 and indicatingultrasonic operation or electrical operation), a desiredperiod/frequency (Δt_(new)) of the control signals (CTRL₁ and CTRL₂)that are used to control the gate drive circuit 25. This desiredperiod/frequency is passed from the power controller 85 to the controlsignal generator 95, which changes the control signals CTRL₁ and CTRL₂in order to change the waveform period to match the desired period. TheCTRL control signals may comprise square wave signals having the desiredperiod or they may comprise periodic pulses with the periodcorresponding to the desired period (Δt_(new)) and with the relativetiming of the pulses of the control signals being set to minimiseharmonic content of the waveform that is generated by the bridge signalgenerator 27 (such as to minimise the 3^(rd) order harmonic). In thisembodiment, the control signals CTRL₁ are output to the FET gate drive37-1 (shown in FIG. 2), which amplifies the control signals and thenapplies them to the FETs 41-1 and 41-2; and the control signals CTRL₂are output to the FET gate drive 37-2 (shown in FIG. 2), which amplifiesthe control signals and then applies them to the FETs 41-3 and 41-4, tothereby generate the desired waveform with the new period (Δt_(new)).

In order to drive the circuitry with the optimal RMS waveform, theMOSFETs 41 are driven as complementary, opposing pairs. Although themaximum output voltage is achieved when the MOSFET pairs are driven at aphase shift of 180 degrees, the resulting harmonic content of such adrive waveform, particularly the 3rd harmonic (which is poorly excludedby the output filter) is quite high. The inventors have found that theoptimal phase shift between the control signals applied to the two pairsof MOSFETs 41, for 3rd harmonic reduction, is around 120°. This isillustrated in FIG. 6 which shows in the upper plot the output from thefirst MOSFET pair 41-1 and 41-2; in the middle plot the output from thesecond MOSFET pair 41-3 and 41-4 (shifted by 120° relative to the upperplot); and in the lower plot the resulting (normalised) output voltageapplied across inputs A and B. The shape of this normalised outputvoltage has very low 3^(rd) order harmonic content.

I & Q Signal Sampling

Both the load voltage and the load current will be substantiallysinusoidal waveforms, although they may be out of phase, depending onthe impedance of the load represented by the transducer 8 and/or thevessel/tissue gripped by the forceps 9. The load current and the loadvoltage will be at the same drive frequency (f_(d)) corresponding to thepresently defined waveform period (Δt_(new)). Normally, when sampling asignal, the sampling circuitry operates asynchronously with respect tothe frequency of the signal that is being sampled. However, as themicroprocessor 23 knows the frequency and phase of the switchingsignals, the synchronous sampling circuit 81 can sample the measuredvoltage/current signal at predefined points in time during the driveperiod. In this embodiment, during the ultrasonic mode of operation, thesynchronous sampling circuit 81 oversamples the measured signal eighttimes per period to obtain four I samples and four Q samples.Oversampling allows for a reduction of errors caused by harmonicdistortion and therefore allows for the more accurate determination ofthe measured current and voltage values. However, oversampling is notessential and indeed under sampling (less than two samples per period)is performed when the device is operating in the electrical mode ofoperation and is possible due to the synchronous nature of the samplingoperation. The timing that the synchronous sampling circuit 81 makesthese samples is controlled, in this embodiment, by the control signalsCTRL₁ and CTRL₂. Thus when the period of these control signals ischanged, the period of the sampling control signals CTRL₁ and CTRL₂ alsochanges (whilst their relative phases stay the same). In this way, thesampling circuitry 81 continuously changes the timing at which itsamples the sensed voltage and current signals as the period of thedrive waveform is changed so that the samples are always taken at thesame time points within the period of the drive waveform. Therefore, thesampling circuit 81 is performing a “synchronous” sampling operationinstead of a more conventional sampling operation that just samples theinput signal at a fixed sampling rate defined by a fixed sampling clock.Of course, such a conventional sampling operation could be used instead.

Measurements

The samples obtained by the synchronous sampling circuitry 51 are passedto the measured voltage and current processing module 83 which candetermine the magnitude and phase of the measured signal from just one“I” sample and one “Q” sample of the load current and load voltage.However, in this embodiment, to achieve some averaging, the processingmodule 83 averages consecutive “I” samples to provide an average “I”value and consecutive “Q” samples to provide an average “Q” value; andthen uses the average I and Q values to determine the magnitude andphase of the measured signal. Of course, it should be recognised thatsome pre-processing of the data may be required to convert the actualmeasured I and Q samples into I and Q samples of the load voltage or theload current, for example, scaling, integration or differentiation ofthe sample values may be performed to convert the sampled values intotrue samples of the load voltage (V_(L)) and the load current (i_(L)).Where integration or differentiation is required, this can be achievedsimply by swapping the order of the I and Q samples—asintegrating/differentiating a sinusoidal signal simply involves ascaling and a 90 degree phase shift.

The RMS load voltage, the RMS load current and the delivered power,P_(delivered), can then be determined from:

$V_{RMS} = {\frac{1}{\sqrt{2}}\sqrt{\left( {V_{I}^{2} + V_{Q}^{2}} \right)}}$$I_{RMS} = {\frac{1}{\sqrt{2}}\sqrt{\left( {I_{I}^{2} + I_{Q}^{2}} \right)}}$${Power} = {{V \cdot I^{*}} = {{\frac{1}{\sqrt{2}}\left( {V_{I} + {jV_{Q}}} \right)\left( {I_{I} - {jI}_{Q}} \right)} = {P_{delivered} + {jP_{reactive}}}}}$$P_{delivered} = {\frac{1}{\sqrt{2}}\left( {{V_{I}I_{I}} + {V_{Q}I_{Q}}} \right)}$$P_{reactive} = {\frac{1}{\sqrt{2}}\left( {{V_{Q}I_{I}} - {V_{I}I_{Q}}} \right)}$Power = V_(RMS)I_(RMS) = P_(delivered) + jP_(reactive)

The impedance of the load represented by the ultrasonic transducer 8 andthe vessel/tissue gripped by the forceps 9 (or just the impedance of theforceps 9 and the vessel/tissue gripped by the forceps 9 if theelectrical energy is directly applied to the forceps 9) can bedetermined from:

$Z_{Load} = {\frac{\left( {V_{I} + {jV_{Q}}} \right)}{\left( {I_{I} + {jI}_{Q}} \right)} = {\frac{\left( {V_{I} + {jV}_{Q}} \right)\left( {I_{I} - {jI}_{Q}} \right)}{\left( {I_{I} + {jI}_{Q}} \right)\left( {I_{I} - {jI}_{Q}} \right)} = {\frac{\left( {{V_{I}I_{I}} + {V_{Q}I_{Q}} + {{jV}_{Q}I_{I}} - {{jjV}_{I}I_{Q}}} \right)}{\sqrt{2}I_{RMS}^{2}} = {R_{Load} + {jX}_{Load}}}}}$

An alternative way of computing R_(Load) and X_(Load) is as follows:

$R_{Load} = {{\frac{P_{delivered}}{\sqrt{2}I_{RMS}^{2}}\mspace{14mu} X_{Load}} = \frac{P_{reactive}}{\sqrt{2}I_{RMS}^{2}}}$

and the phase difference between the load voltage and the load currentcan be determined from:

Phase_(measured) =a tan 2(P _(reactive) ,P _(delivered))

A computationally efficient, approximation to the a tan 2 function canbe made using look up tables and interpolation in fixed pointarithmetic, or using a ‘CORDIC’ like algorithm,

Limits

As with any system, there are certain limits that can be placed on thepower, current and voltage that can be delivered either to theultrasonic transducer 8 or to the forceps 9. The limits used in thisembodiment and how they are controlled will now be described.

In this embodiment, the drive circuitry 29 is designed to deliverultrasonic energy into tissue or to deliver electrical energy intotissue with the following requirements:

-   1) Supplied with a nominally 14V DC supply-   2) Substantially sinusoidal output waveform at approximately 50 kHz    in the case of ultrasonic operation-   3) Substantially sinusoidal output waveform at approximately 500 kHz    in the case of RF electrical operation-   4) Power limited output of 90 W in the case of ultrasonic operation-   5) Power limited output of 100 W in the case of electrical operation-   6) Current limited to 1.4A_(rms) and voltage limited to 130V_(rms)    in the case of ultrasonic operation-   7) Current limited to 1.4A_(rms) and voltage limited to 100V_(rms)    in the case of electrical operation-   8) In the case of ultrasonic operation, the measured phase is    greater than a system defined phase limit

The power controller 85 maintains data defining these limits and usesthem to control the decision about whether to increase or decrease thewaveform period or whether to skip pulses of the control signals giventhe latest measured power, load impedance and/or measured phase. In thisembodiment, when operating in the ultrasonic mode of operation, thephase limit that is used depends on the measured load impedance. Inparticular, the power controller 85 maintains a look up table (notshown) relating load impedance to the phase limit; and the values inthis table limit the phase so that when the measured load impedance islow (indicating that the jaws of the forceps 9 are open and not grippingtissue or a vessel), the delivered power is reduced (preferably tozero).

As discussed above, one of the ways to control the operation of thedevice (when operating in the electrical mode of operation) is tomaximise the drive efficiency. When controlling the device in this way,the power controller 85 tracks a maximum power delivery condition as theload changes. The way that this can be done will now be described.

Maximum Power Delivery Tracking Condition

The complex impedance of the circuitry shown in FIG. 3 (when operatingin the electrical mode of operation and with inductor 48 removed) can beapproximated by the following equation:

$Z = {{j\; 2\; \pi \; {fL}_{S}^{F}} + \frac{1}{J\; 2\; \pi \; {fC}_{S}^{F}} + \frac{j\; 2\; \pi \; {fL}_{M}^{F}R_{{load}\_ {ref}}}{{j\; 2\; \pi \; {fL}_{M}^{F}} + R_{{load}\_ {ref}}} + R_{s}}$

Where:

R_(load_ref) is the load resistance referred to the primary (by thesquare of the turns ratio); and R_(s) represents the equivalent seriesresistance of the inductor, transformer capacitor and switching devices.This complex impedance may be rewritten as:

$Z = {{j\; 2\; \pi \; {fL}_{S}^{F}} + \frac{1}{J\; 2\; \pi \; {fC}_{S}^{F}} + {\frac{4\pi^{2}f^{2}L_{M}^{F^{2}}{\, R_{{load}\_ {ref}}}}{4\pi^{2}f^{2}L_{M}^{F^{2}}{\,{+ R_{{load}\_ {ref}}^{2}}}}\frac{j\; 2\; \pi \; {fL}_{M}^{F}{\, R_{{load}\_ {ref}}^{2}}}{4\pi^{2}f^{2}L_{M}^{F^{2}}{\,{+ R_{{load}\_ {ref}}^{2}}}}} + R_{s}}$

Therefore, the real part of this complex impedance is:

${(Z)} = {\frac{4\pi^{2}f^{2}L_{M}^{F\; 2}R_{{load}\_ {ref}}}{{4\pi^{2}f^{2}L_{M}^{F\; 2}} + R_{{load}\_ {ref}}^{2}} + R_{s}}$

And the imaginary part of this complex impedance is:

${(Z)} = {{2\pi \; {fL}_{S}^{F}} - \frac{1}{2\pi \; {fC}_{S}^{F}} + \frac{2\pi \; {fL}_{M}^{F}R_{{load}\_ {ref}}^{2}}{{4\pi^{2}f^{2}L_{M}^{F\; 2}} + R_{{load}\_ {ref}}^{2}}}$

When the drive frequency (t) corresponds to the resonant frequency ofthis complex impedance, the imaginary part

(Z)=0. Therefore, the power controller 85 can vary the drive frequency(t) to keep the imaginary part

(Z) at or around zero using a phase locked loop. Indeed, it can be shownthat when

(Z)=0 the maximum power (for a given supply voltage) is delivered to theload.

FIG. 7 is a contour plot showing the power contours than can bedelivered to the load versus the drive frequency and the load resistance(Rload). As shown in FIG. 7, the power that can be delivered varies withthe load resistance and the drive frequency. FIG. 7 also shows the line92 of maximum power delivery that can be achieved as the load resistanceand drive frequency change. Therefore, the power controller 85 can usethe measured value of Rload together with stored data defining the line92 shown in FIG. 7 (which may be a look-up-table) to determine thecorresponding drive frequency to be used. In this way, themicroprocessor 23 will track along the line 92 shown in FIG. 7 as theload resistance changes during the cutting/cauterisation process.

One of the advantages of this approach is that it enables a usefuloperating condition at low values of Rload, in particular for values ofRload less than the critical value (i.e. when in which a maximum powerwill be delivered even if this is below the R_(load_ref)<2π∫L^(F) _(M)),desired power level. However, operating along the line 92 of maximumpower delivery can result in some of the above system limits beingbreached unless further control action is taken. In the preferredembodiment, this further control action is to use pulse skippingtechniques until the average power, current and/or voltage is below therelevant limit. For example, as can be seen from the measurementsdescribed above, the measured voltage and current processing module 83can determine the delivered power, the RMS voltage and the RMS current.The power controller 85 can therefore use these values to skip one ormore pulses of the CTRL control signals until the measured voltage andcurrent values are below the relevant system limits and the deliveredpower is at or below the power set-point defined by the medical devicecontrol module 89.

Pulses may be skipped, for example, by passing the pulses generated bythe control signal generator 87 through a logic gate (not shown) andselectively suppressing pulses that are generated by the control signalgenerator 87 by controlling the logic level of another input to thelogic gate. For example, the pulses of each control signal that isgenerated by the control signal generator 87 may be passed through anAND gate, with another input of the AND gate being generated by thepower controller 85 and being a logic “1” when the pulses are to beoutput to the FET gate drives 37 as normal and being a logic “0” whenthe pulses are to be skipped or suppressed. Other pulse skippingtechniques could of course be used.

Medical Device Control Module

As mentioned above, the medical device control module 89 controls thegeneral operation of the cauterisation/cutting device 1. It receivesuser inputs via the user input module 91. These inputs may specify thatthe jaws of the forceps 9 are now gripping a vessel or tissue and thatthe user wishes to begin cutting/cauterisation and specify whetherultrasonic energy or electrical energy is to be applied to thevessel/tissue. In response, in this embodiment, the medical devicecontrol module 89 initiates a cutting/cauterisation control procedure.Initially, the medical device control module 89 sends an initiationsignal to the power controller 85 and obtains the load impedancemeasurements determined by the measured voltage and current processingmodule 83. The medical device control module 89 then checks the obtainedload impedance to make sure that the load is not open circuit or shortcircuit. If it is not, then the medical device control module 89 startsto vary the power set point to perform the desired cutting/cauterisationand sets the initial period/frequency of the drive signal to begenerated. As discussed above, for ultrasonic operation, the initialfrequency of the drive signal will be set around 50 kHz and for RFelectrical operation, the initial frequency will be set around 500 kHz.

Voltage/Current Sensing Circuitry

As shown in FIG. 2, voltage sensing circuitry 31 is provided to sensethe load voltage applied to the load and current sensing circuitry 33 isprovided to sense the current applied to the load. The sensed signalsare supplied to the microprocessor 23 for use in controlling theoperation of the medical device. There are various ways of sensing theload voltage and the load current and some of these will now bedescribed.

FIG. 8a illustrates the primary side of the first transformer circuit42-1 and one way in which the current sensing circuitry 33 obtains ameasure of the load current. As shown, the current sensing circuitry 33comprises an additional inductor turn(s) 67 which link the flux presentin inductor 47 (or inductor 49) and that consequently outputs a voltageacross inductor 67 that varies with the rate of change of load current.The voltage across the inductor 67 is a bipolar voltage whose amplitudeis directly proportional to the rate of change of load current and thenumber of turns in 67. This bipolar voltage is scaled and converted intoa unipolar voltage suitable for input to the microprocessor 23 by theop-amp circuit 69-1 which outputs a measured voltage (V^(meas)). Thismeasured voltage will also depend on the current flowing in the inductor47 and so will also depend on the current flowing on the secondary sideof the transformer circuit 42-1 and thus the current flowing through theload. As the ratio of the number of turns of inductor 47 to inductor 67is known, the measured voltage and current processing module 83 can useV^(meas) to determine the voltage across inductor 47. The voltage acrossinductor 47 is related to the current flowing through the inductor 47 byV=Ldi/dt. As the inductance of inductor 47 is known, the measuredvoltage and current processing module 83 can determine the currentflowing in the primary side of the transformer circuit 42-1 byintegrating the voltage across the inductor 47 and by scaling the resultto account for the inductance of the inductor 47 (and the scaling of theop-amp-circuit 69-1). This current measure can then be converted into asuitable measure of the load current (i_(L)) by a further scaling totake into account the number of turns between inductor 49 and inductor53. Of course, the measured voltage and current processing module 83does not need to integrate the voltage across the inductor 47—as themeasured signals are sinusoidal and so integration can be achieved byapplying a suitable scaling factor and a 90 degree phase shift. Thus,the measured voltage and current processing module 83 can determine theload current by applying a suitable (pre-stored) scale factor to themeasured voltage (V^(meas)) and by applying a suitable 90 degree phaseshift (which can be achieved simply by swapping the order of the I and Qsamples as discussed above).

FIG. 8b illustrates the primary side of the first transformer circuit42-1 and another way in which the current sensing circuitry 33 canobtain a measure of the load current. As shown, in this case, thecurrent sensing circuitry 33 measures the voltage across the capacitor45. The voltage across the capacitor 45 is a bipolar voltage. Thisbipolar voltage is scaled and converted into a unipolar voltage suitablefor input to the microprocessor 23 by the op-amp circuit 69-2 whichoutputs a measured voltage (V^(meas)). This voltage is related to thecurrent flowing in the primary side of the transformer circuit 42 byI=CdV^(meas)/dt and thus the current flowing through the load. As thecapacitance of the capacitor 45 is known, the measured voltage andcurrent processing module 83 can determine the current flowing in theprimary side of the transformer circuit 42-1 by differentiating thevoltage across the capacitor 45 and by scaling the result to account forthe capacitance of the capacitor 45 (and the scaling of theop-amp-circuit 69-2). This current measure can then be converted into asuitable measure of the load current (i_(L)) by a further scaling totake into account the number of turns between inductor 49 and inductor53. Of course, the measured voltage and current processing module 83does not need to differentiate the voltage across the capacitor 45—asthe measured signals are sinusoidal and so differentiation can beachieved by applying a suitable scaling factor and a 90 degree phaseshift. Thus, the measured voltage and current processing module 83 candetermine the load current by applying a suitable (pre-stored) scalefactor to the measured voltage (V^(meas)) and by applying a suitable 90degree phase shift (which can be achieved simply by swapping the orderof the I and Q samples as discussed above).

FIG. 8c schematically illustrates how a measure of the load voltage canbe determined. FIG. 8c shows the secondary side of the first transformercircuit 42 and illustrates the use of a voltage divider circuit (in thiscase formed by resistors R1 and R2), with the voltage across resistor R2being input to the op-amp circuit 69-3. Therefore, by applying anappropriate scale on the measured voltage from op-amp 69-3, the measuredvoltage and current processing module 83 can determine the load voltage.

The sensing circuits described above use op-amp circuits 69 to convertthe bipolar drive signals into unipolar voltages that are suitable forinput to the microprocessor 23. The use of such op-amp circuits has anumber of disadvantages, including that they are costly, they consumepower and they requires space within the electronics. These areimportant factors when the device is designed to be battery powered andthe electronics are housed within the handle 5 of the device. FIG. 9illustrates various sensing circuits that can be used without an op-amp.The circuit of FIG. 9a is suitable when the AC drive signal is unipolar.This may be achieved by replacing the full bridge signal generator 27with a half bridge signal generator. This would mean removing, forexample, the FETs 41-3 and 41-4 and connecting node B to ground. In thiscase, the microprocessor 23 only needs to generate one control signal(CTRL₁) to control FETs 41-1 and 41-2. The circuit of FIG. 9b issuitable for both unipolar and bipolar drive signals.

In FIG. 9a , the capacitor 45 has been moved to be between the inductor49 that forms the primary of the transformer circuit 42-1 and ground(GND). The current sensing circuitry 33 is arranged to measure thevoltage across the capacitor 45 via a potential divider formed byresistors R1//R2 and R3. The resistor R1 connects the output of a DCblocking capacitor C_(B) to a supply voltage rail of the microprocessor23 (in this case at 3.3V) through the switch 121; and resistor R2connects the output of the DC blocking capacitor C_(B) to a referencepotential (in this case ground). The resistors R1 and R2 thereforeprovide a divider circuit that applies a DC bias to the measured ACsignal. The DC blocking capacitor prevents this DC bias from coupling tothe drive circuit. Typically, the resistors R1 and R2 are equal so thatthe DC bias will be at 1.65V. Thus, the voltage output from the sensingcircuit 33 will be an AC voltage having a mid-rail value of about 1.65 Vand whose peak voltage will be a proportion of the voltage across thecapacitor 45. The potential divider formed by resistors R1//R2 and R3 issuch that the peak to peak amplitude of the AC signal passed to themicroprocessor 23 is less than the 3.3V input range of themicroprocessor 23. If the microprocessor 23 operates at a differentvoltage rail (for example 5V), then the values of the resistors R1, R2and R3 can be adjusted accordingly. To minimise current drawn eitherfrom the 3.3V rail or from the transformer circuit 42-1, the resistorsR1, R2 and R3 can have relatively large values and typical values forthese resistors are: R1=R2=200Ω and R3=1000Ω. The values of R1 and R2should be chosen to meet the input impedance requirements of theAnalogue to digital converter used to sample the signals. The switch 121allows the microprocessor 23 to disconnect the sensing circuit 33 fromthe 3.3V rail—so that when sensing is not required, the circuit 33 doesnot consume any power.

FIG. 9b illustrates the way in which similar circuits can be provided onthe secondary side of the transformer circuit 42-1. In particular, FIG.9b shows the voltage sensing circuit 31 used to obtain a measure of theload voltage V_(L) via a voltage divider formed by capacitors C1 and C2.If the capacitors C1 and C2 were replaced with resistors, then ablocking capacitor C_(B) should be provided before the divider circuitformed by resistors R1 and R2 in the manner shown in FIG. 9a . FIG. 9balso shows the current sensing circuit 33, which senses the load currentby sensing the voltage across capacitor C3 via a potential dividerformed by resistors R4//R5 and R6. As shown, a DC blocking capacitorC_(B) is provided to enable a DC bias signal to be added to the outputto the microprocessor via the divider circuit formed from resistors R4and R5. As before, the switches 121-1 and 121-2 allow the microprocessorto switch off the sensing circuits 31 and 33 when sensor signals are notrequired. As those skilled in the art will appreciate, the “op-ampless”sensing circuits 31 and 33 illustrated in FIG. 9 are cheaper tomanufacture, can be made to consume less power and take up less space onthe circuit board than the op-amp circuits illustrated in FIG. 8.

The circuits illustrated in FIGS. 8 and 9 were used to obtainmeasurements from the first transformer circuit 42-1 of the drivecircuit 29. As those skilled in the art will appreciate, the same orsimilar sensing circuits would be provided for sensing signals in thesecond transformer circuit 42-2. Further, although the sensing circuitsillustrated in FIG. 9 sense the voltage across a capacitor, the circuitscould also sense the voltage across another impedance element, such asacross a resistor of the transformer circuit 42.

Active Battery Protection

Typically, with battery operated devices such as the medical device 1described above, a fuse is provided between the batteries and theelectric circuitry, to protect the battery from damage caused by shortcircuits and the like. However, standard fuses have a resistance ofabout 10 mOhms. With such a standard fuse, when 10A is drawn from thebatteries, approximately 1 W is dissipated through the fuse. An activefuse circuit 130 that is used in this embodiment will now be describedwith reference to FIG. 10 that reduces the power dissipation associatedwith such standard fuses.

FIG. 10 shows the batteries 15, which supply the 14V rail and the GNDrail to the circuitry shown in FIG. 3. The active fuse circuit 130comprises a differential amplifier 131 that measures the voltage dropacross part of a PCB conductor trace 133 that is connected between the14V rail and the positive terminal of the batteries (V_(bat+)). Theconductor trace 133 has a resistance of approximately 1-2 mOhms and sothe voltage drop is proportional to the current that is being drawn fromthe batteries 15. The measured voltage drop is low pass filtered by thelow pass filter 135 to avoid transient spikes triggering the fusecircuit—so that it is just the voltage corresponding to the DC currentdrawn from the batteries 15 that will pass through the filter 135. Inthis embodiment, the low pass filter 135 has a cut-off frequency ofabout 10 Hz. The output from the low pass filter 135 is then comparedwith a reference voltage (V_(ref)) using a latching comparator 137. Thereference voltage is set in advance to correspond to a desired limit onthe current drawn from the batteries. In this embodiment, V_(ref) is setto correspond to a current limit of 15A. When the voltage drop acrossthe trace 133 is less than the reference voltage the output of thecomparator 137 remains at a high value—which maintains the FET switch139 on and so current can be drawn from the batteries 15 by the bridgesignal generator 27. However, when the voltage drop across the trace 133is greater than the reference voltage, then the output of the comparator137 goes low and it is maintained low even if the current drawn from thebatteries drops below the defined limit. When the comparator output islow the FET 139 is switched off, thereby disconnecting the batteries 15from at least the bridge signal generator 27.

In this embodiment, the FET 139 is an N-channel enhanced mode switchhaving an on resistance of just 2 mOhms. This means that when the switch139 is switched on and 10A is being drawn from the batteries, just 0.2 Wis dissipated through the switch 139.

In this embodiment, when the comparator 137 is triggered and the switch139 is switched off (open circuit), the batteries 15 have to be removedto reset the active fuse circuitry 130—as the opening of the switch 139disconnects all the electronics except the circuit components of theactive fuse 130 from the batteries. Alternatively, if the microprocessor23 (or some other control circuitry) is powered directly from thebatteries, then the comparator 137 could be reset either in response toa user input (for example in response to the user pressing a resetbutton or the like) or in response to some other trigger event (such asafter a predetermined time-out period).

Modifications and Alternatives

A medical cauterisation/cutting device has been described above. Asthose skilled in the art will appreciate, various modifications can bemade and some of these will now be described. Other modifications willbe apparent to those skilled in the art.

In the above embodiment, various operating frequencies, currents,voltages etc were described. As those skilled in the art willappreciate, the exact currents, voltages, frequencies, capacitor values,inductor values etc. can all be varied depending on the application andany values described above should not be considered as limiting in anyway. However, in general terms, the circuit described above has beendesigned to provide a drive signal to a medical device, where thedelivered power is desired to be at least 10 W and preferably between 10W and 200 W; the delivered voltage is desired to be at least 20 V_(Rms)and preferably between 30 V_(Rms) and 120 V_(Rms); the delivered currentis designed to be at least 0.5 A_(RMS) and preferably between 1 A_(RMS)and 2 A_(RMS); and the drive frequency for ultrasonic operation isdesired to be at least 20 kHz and preferably between 30 kHz and 80 kHz;and the drive frequency for RF operation is desired to be at least 100kHz and preferably between 250 kHz and 1 MHz.

In the above embodiment, the resonant circuits 43-1 and 43-2 were formedfrom capacitor-inductor-inductor elements. As those skilled in the artwill appreciate, other resonant circuit designs with multiple capacitorsand inductors in various series and parallel configurations or simplerLC resonant circuits may also be used. Also, in some applications thereis no need for a transformer to step-up the drive voltage, as the FETscan deliver the required drive voltage.

FIG. 1 illustrates one way in which the batteries and the controlelectronics can be mounted within the handle of the medical device. Asthose skilled in the art will appreciate, the form factor of the handlemay take many different designs. Indeed, it is not essential for thedevice to be battery powered, although this is preferred for someapplications to avoid the need for power cords and the like.

The embodiment described above included a description of various novelfeatures, including the novel ability to selectively apply ultrasonicenergy or RF energy to the tissue gripped by the forceps, the novel wayin which the microprocessor controlled the operation of the device inthe electrical mode of operation; the way in which load current/voltageis measured and the way in which the batteries are protected using anactive fuse circuit. As those skilled in the art will appreciate, thesenovel features do not need to be employed together. For example, thecurrent/voltage sensing techniques described above can be used withother devices as can the active fuse circuit. Similarly, the way inwhich the electrical mode of operation is controlled by tracking themaximum power delivery condition and by using pulse skipping techniquescan be used in a device that does not have an ultrasonic transducer.

In the above embodiment, an exemplary control algorithm for performingthe cutting/cauterisation of the vessel or tissue gripped by the forcepswas described. As those skilled in the art will appreciate, variousdifferent procedures may be used and the reader is referred to theliterature describing the operation of such cutting/cauterisationdevices for further details.

In the above embodiment, four FET switches were used to convert the DCvoltage provided by the batteries into an alternating signal at thedesired frequency. As those skilled in the art will appreciate, it isnot necessary to use four switches—two switches may be used instead(using a half bridge circuit). Additionally, although FET switches wereused, other switching devices, such as bipolar transistor switches maybe used instead. However, MOSFETs are preferred due to their superiorperformance in terms of low losses when operating at the above describedfrequencies and current levels.

In the above embodiment, the I & Q sampling circuitry 81 oversampled thesensed voltage/current signal in the ultrasonic mode of operation andundersampled the sensed voltage/current signal in the electrical mode ofoperation. As those skilled in the art will appreciate, this is notessential. Because of the synchronous nature of the sampling, samplesmay be taken more than once per period or once every n^(th) period ifdesired. The sampling rate used in the above embodiment was chosen tomaximise the rate at which measurements were made available to the powercontroller 85 and the medical device control module 89 as this allowsfor better control of the applied power during the cauterisationprocess.

In the above embodiment, a 14V DC supply was provided. In otherembodiments, lower (or higher) DC voltage sources may be provided. Inthis case, a larger (or smaller) transformer turns ratio may be providedto increase the load voltage to the desired level or lower operatingvoltages may be used.

In the above embodiment, the medical device was arranged to deliver adesired power (in the form of ultrasonic energy or electrical energy) tothe tissue/vessel gripped by the forceps. In an alternative embodiment,the device may be arranged to deliver a desired current or a desiredvoltage level to the ultrasonic transducer or the forceps.

In the above embodiment the battery is shown integral to the medicaldevice. In an alternative embodiment the battery may be packaged so asto clip on a belt on the surgeon or simply be placed on the Mayo stand.In this embodiment a relatively small two conductor cable would connectthe battery pack to the medical device.

In the above embodiment, a microprocessor based control circuitry wasprovided. This is preferred due to the ease with which themicroprocessor can be programmed to perform the above control actionsusing appropriate computer software. Such software can be provided on atangible carrier, such as a CD-ROM or the like. Alternatively, hardwarecontrol circuitry can be used in place of the microprocessor basedcircuitry described above.

In the above embodiment, the user controlled whether the energydelivered to the vessel/tissue was ultrasonic energy or RF electricalenergy. In alternative embodiments, the microprocessor may control theselection based on an internally generated control signal or in responseto a control signal received from another device.

In the above embodiment, the active fuse circuit opened a switch thatdisconnected the bridge signal generator from the batteries. In analternative embodiment, the active fuse circuit could instead switch ina large impedance between the batteries and the bridge signal generatorto limit the current drawn from the batteries. Also, the switch could beused to disconnect the positive voltage supply from the signal generatorinstead of disconnecting the negative terminal of the batteries.

1.-17. (canceled)
 18. A medical device comprising: an end effector for gripping a vessel/tissue; a drive circuit for generating a cyclically varying drive signal for driving energy into the vessel/tissue; sensing circuitry for sensing a drive signal generated by the drive circuit; and a controller responsive to the sensing circuitry and operable to control the drive circuit to control the energy delivered to the vessel/tissue; wherein the drive circuit comprises an impedance element that is coupled to a reference potential and wherein the sensing circuitry comprises a divider circuit for obtaining a measure of the voltage across the impedance element and a bias signal generator for applying a DC bias signal to the voltage measure.
 19. A device according to claim 18, wherein the impedance element comprises a capacitor or a resistor.
 20. A device according to claim 18, wherein the sensing circuitry comprises DC blocking circuitry for preventing the DC bias signal from the bias signal generator from coupling with the drive circuit.
 21. A device according to claim 18, wherein the bias signal generator comprises a divider circuit connected between a reference voltage and a supply voltage of the controller.
 22. A device according to claim 21, wherein the divider circuit of the bias signal generator is connected to the supply voltage of the controller via a switch and wherein the controller is configured to open the switch when the controller does not require signals from the sensing circuitry. 23.-39. (canceled) 